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Power Factor Controllers
The MC34262/MC33262 are active power factor controllers specifically designed for use as a preconverter in electronic ballast and in off-line power converter applications. These integrated circuits feature an internal startup timer for stand-alone applications, a one quadrant multiplier for near unity power factor, zero current detector to ensure critical conduction operation, transconductance error amplifier, quickstart circuit for enhanced startup, trimmed internal bandgap reference, current sensing comparator, and a totem pole output ideally suited for driving a power MOSFET. Also included are protective features consisting of an overvoltage comparator to eliminate runaway output voltage due to load removal, input undervoltage lockout with hysteresis, cycle-by-cycle current limiting, multiplier output clamp that limits maximum peak switch current, an RS latch for single pulse metering, and a drive output high state clamp for MOSFET gate protection. These devices are available in dual-in-line and surface mount plastic packages. * Overvoltage Comparator Eliminates Runaway Output Voltage
MC34262 MC33262
POWER FACTOR CONTROLLERS
SEMICONDUCTOR TECHNICAL DATA
8 1
P SUFFIX PLASTIC PACKAGE CASE 626
* * * * * * * *
Internal Startup Timer One Quadrant Multiplier Zero Current Detector Trimmed 2% Internal Bandgap Reference Totem Pole Output with High State Clamp Undervoltage Lockout with 6.0 V of Hysteresis Low Startup and Operating Current Supersedes Functionality of SG3561 and TDA4817
8
1
D SUFFIX PLASTIC PACKAGE CASE 751 (SO-8)
PIN CONNECTIONS Simplified Block Diagram
Zero Current Detector 5 Zero Current Detect Input Voltage Feedback Input Compensation Multiplier Input Current Sense Input 1 2 3 4 (Top View) 8 VCC 7 Drive Output 6 Gnd 5 Zero Current Detect Input
2.5V Reference
Undervoltage Lockout
VCC 8
Drive Output Multiplier, Latch, PWM, Timer, & Logic 7 Current Sense Input
Overvoltage Comparator + Error Amp 1.08 Vref
4
ORDERING INFORMATION
Device Operating Temperature Range TA = 0 to +85C TA = - 40 to +105C Package SO-8 Plastic DIP SO-8 Plastic DIP
Rev 1
Multiplier Input 3
+
Multiplier Quickstart Gnd 6 Compensation 2
Vref
Voltage Feedback 1 Input
MC34262D MC34262P MC33262D MC33262P
(c) Motorola, Inc. 1996
MOTOROLA ANALOG IC DEVICE DATA
1
MC34262 MC33262
MAXIMUM RATINGS
Rating Total Power Supply and Zener Current Output Current, Source or Sink (Note 1) Current Sense, Multiplier, and Voltage Feedback Inputs Zero Current Detect Input High State Forward Current Low State Reverse Current Power Dissipation and Thermal Characteristics P Suffix, Plastic Package, Case 626 Maximum Power Dissipation @ TA = 70C Thermal Resistance, Junction-to-Air D Suffix, Plastic Package, Case 751 Maximum Power Dissipation @ TA = 70C Thermal Resistance, Junction-to-Air Operating Junction Temperature Operating Ambient Temperature (Note 3) MC34262 MC33262 Storage Temperature Symbol (ICC + IZ) IO Vin Iin 50 -10 Value 30 500 -1.0 to +10 Unit mA mA V mA
PD RJA PD RJA TJ TA
800 100 450 178 +150 0 to + 85 - 40 to +105
mW C/W mW C/W C C
Tstg
- 65 to +150
C
ELECTRICAL CHARACTERISTICS (VCC = 12 V (Note 2), for typical values TA = 25C, for min/max values TA is the operating ambient temperature range that applies [Note 3], unless otherwise noted.)
Characteristic ERROR AMPLIFIER Voltage Feedback Input Threshold TA = 25C TA = Tlow to Thigh (VCC = 12 V to 28 V) Line Regulation (VCC = 12 V to 28 V, TA = 25C) Input Bias Current (VFB = 0 V) Transconductance (TA = 25C) Output Current Source (VFB = 2.3 V) Sink (VFB = 2.7 V) Output Voltage Swing High State (VFB = 2.3 V) Low State (VFB = 2.7 V) OVERVOLTAGE COMPARATOR Voltage Feedback Input Threshold MULTIPLIER Input Bias Current, Pin 3 (VFB = 0 V) Input Threshold, Pin 2 Dynamic Input Voltage Range Multiplier Input (Pin 3) Compensation (Pin 2) Multiplier Gain (VPin 3 = 0.5 V, VPin 2 = Vth(M) + 1.0 V) (Note 4) ZERO CURRENT DETECTOR Input Threshold Voltage (Vin Increasing) Hysteresis (Vin Decreasing) Input Clamp Voltage High State (IDET = + 3.0 mA) Low State (IDET = - 3.0 mA) Vth VH VIH VIL 1.33 100 6.1 0.3 1.6 200 6.7 0.7 1.87 300 -- 1.0 V mV V IIB Vth(M) VPin 3 VPin 2 K -- 1.05 VOL(EA) 0 to 2.5 Vth(M) to (Vth(M) + 1.0) 0.43 - 0.1 1.2 VOL(EA) 0 to 3.5 Vth(M) to (Vth(M) + 1.5) 0.65 - 0.5 -- -- -- 0.87 1/V A V V VFB(OV) 1.065 VFB 1.08 VFB 1.095 VFB V VFB 2.465 2.44 Regline IIB gm IO -- -- VOH(ea) VOL(ea) 5.8 -- 10 10 6.4 1.7 -- -- V -- 2.4 -- -- 80 2.5 -- 1.0 - 0.1 100 2.535 2.54 10 - 0.5 130 mV A mho A V Symbol Min Typ Max Unit
2
MOTOROLA ANALOG IC DEVICE DATA
MC34262 MC33262
ELECTRICAL CHARACTERISTICS (VCC = 12 V (Note 2), for typical values TA = 25C, for min/max values TA is the operating ambient temperature range that applies (Note 3), unless otherwise noted.)
Characteristic CURRENT SENSE COMPARATOR Input Bias Current (VPin 4 = 0 V) Input Offset Voltage (VPin 2 = 1.1 V, VPin 3 = 0 V) Maximum Current Sense Input Threshold (Note 5) Delay to Output DRIVE OUTPUT Output Voltage (VCC = 12 V) Low State (ISink = 20 mA) Low State (ISink = 200 mA) High State (ISource = 20 mA) High State (ISource = 200 mA) Output Voltage (VCC = 30 V) High State (ISource = 20 mA, CL = 15 pF) Output Voltage Rise Time (CL = 1.0 nF) Output Voltage Fall Time (CL = 1.0 nF) Output Voltage with UVLO Activated (VCC = 7.0 V, ISink = 1.0 mA) RESTART TIMER Restart Time Delay UNDERVOLTAGE LOCKOUT Startup Threshold (VCC Increasing) Minimum Operating Voltage After Turn-On (VCC Decreasing) Hysteresis TOTAL DEVICE Power Supply Current Startup (VCC = 7.0 V) Operating Dynamic Operating (50 kHz, CL = 1.0 nF) Power Supply Zener Voltage (ICC = 25 mA)
NOTES: 1. Maximum package power dissipation limits must be observed. 2. Adjust VCC above the startup threshold before setting to 12 V. 3. Tlow = 0C for MC34262 3. Tlow = - 40C for MC33262 Thigh = + 85C for MC34262 Thigh = +105C for MC33262
Symbol
Min
Typ
Max
Unit A mV V ns
IIB VIO Vth(max) tPHL(in/out)
-- -- 1.3 --
- 0.15 9.0 1.5 200
-1.0 25 1.8 400
V VOL VOH VO(max) 14 tr tf VO(UVLO) -- -- -- 16 50 50 0.1 18 120 120 0.5 ns ns V -- -- 9.8 7.8 0.3 2.4 10.3 8.4 0.8 3.3 -- -- V
tDLY Vth(on) VShutdown VH ICC
200
620
--
s
11.5 7.0 3.8
13 8.0 5.0
14.5 9.0 6.2
V V V
mA -- -- -- 0.25 6.5 9.0 36
Pin 4 Threshold
0.4 12 20 -- V
VZ
4. K =
30
VPin 3 (VPin 2 - Vth(M)) 5. This parameter is measured with VFB = 0 V, and VPin 3 = 3.0 V.
Figure 1. Current Sense Input Threshold versus Multiplier Input
V CS , CURRENT SENSE PIN 4 THRESHOLD (V) 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 - 0.2 VPin 2 = 2.0 V 0.6 1.4 2.2 3.0 VM, MULTIPLIER PIN 3 INPUT VOLTAGE (V) 3.8 VCC = 12 V TA = 25C V CS , CURRENT SENSE PIN 4 THRESHOLD (V) 1.6 0.08 0.07 0.06 0.05 0.04 0.03 0.02 0.01 0 - 0.12
Figure 2. Current Sense Input Threshold versus Multiplier Input, Expanded View
VPin 2 = 3.5 V VPin 2 = 3.25 V
VCC = 12 V TA = 25C
VPin 2 = 2.0 V
- 0.06 0 0.06 0.12 0.18 VM, MULTIPLIER PIN 3 INPUT VOLTAGE (V)
0.24
MOTOROLA ANALOG IC DEVICE DATA
3
MC34262 MC33262
V FB, VOLTAGE FEEDBACK THRESHOLD CHANGE (mV) V FB(OV) , OVERVOLTAGE INPUT THRESHOLD (%V FB )
Figure 3. Voltage Feedback Input Threshold Change versus Temperature
4.0 VCC = 12 V Pins 1 to 2 0
Figure 4. Overvoltage Comparator Input Threshold versus Temperature
110 VCC = 12 V 109
- 4.0 - 8.0 -12 -16 - 55
108
107
- 25
0
25
50
75
100
125
106 - 55
- 25
0
25
50
75
100
125
TA, AMBIENT TEMPERATURE (C)
TA, AMBIENT TEMPERATURE (C)
Figure 5. Error Amp Transconductance and Phase versus Frequency
120 g , TRANSCONDUCTANCE ( mho) m Phase 100 80 60 40 20 0 3.0 k Transconductance VCC = 12 V VO = 2.5 V to 3.5 V RL = 100 k to 3.0 V CL = 2.0 pF TA = 25C 0 30 60 90 120 150 180 3.0 M
Figure 6. Error Amp Transient Response
O , EXCESS PHASE (DEGREES)
4.00 V VCC = 12 V RL = 100 k CL = 2.0 pF TA = 25C 0.75 V/DIV 5.0 s/DIV
3.25 V
2.50 V
10 k
30 k 100 k 300 k f, FREQUENCY (Hz)
1.0 M
Figure 7. Quickstart Charge Current versus Temperature
Vchg , QUICKSTART CHARGE VOLTAGE (V) 1.80 VCC = 12 V 1.76 800 900 I chg , QUICKSTART CHARGE CURRENT ( A) 800 t DLY, RESTART TIME DELAY ( s)
Figure 8. Restart Timer Delay versus Temperature
VCC = 12 V
700
1.72 Voltage 1.68 Current
700
600
600
500
1.64 - 55
- 25
0 25 50 75 TA, AMBIENT TEMPERATURE (C)
100
500 125
400 - 55
- 25
0
25
50
75
100
125
TA, AMBIENT TEMPERATURE (C)
4
MOTOROLA ANALOG IC DEVICE DATA
MC34262 MC33262
Figure 9. Zero Current Detector Input Threshold Voltage versus Temperature
Vsat , OUTPUT SATURATION VOLTAGE (V) 1.7 V th , THRESHOLD VOLTAGE (V) Upper Threshold (Vin, Increasing) 1.6 VCC = 12 V 0 - 2.0 - 4.0 - 6.0 4.0 2.0 0 Sink Saturation (Load to VCC) Gnd 0 80 160 240 320 IO, OUTPUT LOAD CURRENT (mA) Source Saturation (Load to Ground)
Figure 10. Output Saturation Voltage versus Load Current
VCC VCC = 12 V 80 s Pulsed Load 120 Hz Rate
1.5
1.4 Lower Threshold (Vin, Decreasing) 1.3 - 55 - 25 0 25 50 75 TA, AMBIENT TEMPERATURE (C) 100 125
VO , OUTPUT VOLTAGE
Figure 11. Drive Output Waveform
90% VCC = 12 V CL = 1.0 nF TA = 25C
Figure 12. Drive Output Cross Conduction
5.0 V/DIV 125 VCC = 12 V CL = 15 pF TA = 25C
I CC , SUPPLY CURRENT
10%
100 ns/DIV
100 ns/DIV
Figure 13. Supply Current versus Supply Voltage
16 I CC , SUPPLY CURRENT (mA) VCC , SUPPLY VOLTAGE (V) 14 13 12 11 10 9.0 8.0 40 7.0 - 55
Figure 14. Undervoltage Lockout Thresholds versus Temperature
12
Startup Threshold (VCC Increasing)
8.0 VFB = 0 V Current Sense = 0 V Multiplier = 0 V CL = 1.0 nF f = 50 kHz TA = 25C 0 10 20 30 VCC, SUPPLY VOLTAGE (V)
4.0
Minimum Operating Threshold (VCC Decreasing)
0
- 25
0 25 50 75 TA, AMBIENT TEMPERATURE (C)
100
MOTOROLA ANALOG IC DEVICE DATA
5
100 mA/DIV
MC34262 MC33262
FUNCTIONAL DESCRIPTION
Introduction With the goal of exceeding the requirements of legislation on line-current harmonic content, there is an ever increasing demand for an economical method of obtaining a unity power factor. This data sheet describes a monolithic control IC that was specifically designed for power factor control with minimal external components. It offers the designer a simple, cost-effective solution to obtain the benefits of active power factor correction. Most electronic ballasts and switching power supplies use a bridge rectifier and a bulk storage capacitor to derive raw dc voltage from the utility ac line, Figure 15. Figure 15. Uncorrected Power Factor Circuit
Rectifiers AC Line + Bulk Storage Capacitor Load Converter AC Line Voltage
can be made to appear resistive to the ac line, thus significantly reducing the harmonic current content. Figure 16. Uncorrected Power Factor Input Waveforms
Vpk Rectified DC 0 Line Sag
0 AC Line Current
This simple rectifying circuit draws power from the line when the instantaneous ac voltage exceeds the capacitor voltage. This occurs near the line voltage peak and results in a high charge current spike, Figure 16. Since power is only taken near the line voltage peaks, the resulting spikes of current are extremely nonsinusoidal with a high content of harmonics. This results in a poor power factor condition where the apparent input power is much higher than the real power. Power factor ratios of 0.5 to 0.7 are common. Power factor correction can be achieved with the use of either a passive or an active input circuit. Passive circuits usually contain a combination of large capacitors, inductors, and rectifiers that operate at the ac line frequency. Active circuits incorporate some form of a high frequency switching converter for the power processing, with the boost converter being the most popular topology, Figure 17. Since active input circuits operate at a frequency much higher than that of the ac line, they are smaller, lighter in weight, and more efficient than a passive circuit that yields similar results. With proper control of the preconverter, almost any complex load
The MC34262, MC33262 are high performance, critical conduction, current-mode power factor controllers specifically designed for use in off-line active preconverters. These devices provide the necessary features required to significantly enhance poor power factor loads by keeping the ac line current sinusoidal and in phase with the line voltage. Operating Description The MC34262, MC33262 contain many of the building blocks and protection features that are employed in modern high performance current mode power supply controllers. There are, however, two areas where there is a major difference when compared to popular devices such as the UC3842 series. Referring to the block diagram in Figure 19, note that a multiplier has been added to the current sense loop and that this device does not contain an oscillator. The reasons for these differences will become apparent in the following discussion. A description of each of the functional blocks is given below.
Figure 17. Active Power Factor Correction Preconverter
Rectifiers AC Line + PFC Preconverter Converter
High Frequency Bypass Capacitor
+ MC34362
Bulk Storage Capacitor
Load
6
MOTOROLA ANALOG IC DEVICE DATA
MC34262 MC33262
Error Amplifier An Error Amplifier with access to the inverting input and output is provided. The amplifier is a transconductance type, meaning that it has high output impedance with controlled voltage-to-current gain. The amplifier features a typical gm of 100 mhos (Figure 5). The noninverting input is internally biased at 2.5 V 2.0% and is not pinned out. The output voltage of the power factor converter is typically divided down and monitored by the inverting input. The maximum input bias current is - 0.5 A, which can cause an output voltage error that is equal to the product of the input bias current and the value of the upper divider resistor R2. The Error Amp output is internally connected to the Multiplier and is pinned out (Pin 2) for external loop compensation. Typically, the bandwidth is set below 20 Hz, so that the amplifier's output voltage is relatively constant over a given ac line cycle. In effect, the error amp monitors the average output voltage of the converter over several line cycles. The Error Amp output stage was designed to have a relatively constant transconductance over temperature. This allows the designer to define the compensated bandwidth over the intended operating temperature range. The output stage can sink and source 10 A of current and is capable of swinging from 1.7 V to 6.4 V, assuring that the Multiplier can be driven over its entire dynamic range. A key feature to using a transconductance type amplifier, is that the input is allowed to move independently with respect to the output, since the compensation capacitor is connected to ground. This allows dual usage of of the Voltage Feedback Input pin by the Error Amplifier and by the Overvoltage Comparator. Overvoltage Comparator An Overvoltage Comparator is incorporated to eliminate the possibility of runaway output voltage. This condition can occur during initial startup, sudden load removal, or during output arcing and is the result of the low bandwidth that must be used in the Error Amplifier control loop. The Overvoltage Comparator monitors the peak output voltage of the converter, and when exceeded, immediately terminates MOSFET switching. The comparator threshold is internally set to 1.08 Vref. In order to prevent false tripping during normal operation, the value of the output filter capacitor C3 must be large enough to keep the peak-to-peak ripple less than 16% of the average dc output. The Overvoltage Comparator input to Drive Output turn-off propagation delay is typically 400 ns. A comparison of startup overshoot without and with the Overvoltage Comparator circuit is shown in Figure 23. Multiplier A single quadrant, two input multiplier is the critical element that enables this device to control power factor. The ac full wave rectified haversines are monitored at Pin 3 with respect to ground while the Error Amp output at Pin 2 is monitored with respect to the Voltage Feedback Input threshold. The Multiplier is designed to have an extremely linear transfer curve over a wide dynamic range, 0 V to 3.2 V for Pin 3, and 2.0 V to 3.75 V for Pin 2, Figure 1. The Multiplier output controls the Current Sense Comparator threshold as the ac voltage traverses sinusoidally from zero to peak line, Figure 18. This has the effect of forcing the MOSFET on-time to track the input line voltage, resulting in a fixed Drive Output on-time, thus making the preconverter load appear to be resistive to the ac line. An approximation of the Current Sense Comparator threshold can be calculated from the following equation. This equation is accurate only under the given test condition stated in the electrical table. VCS, Pin 4 Threshold 0.65 (VPin 2 - Vth(M)) VPin 3 A significant reduction in line current distortion can be attained by forcing the preconverter to switch as the ac line voltage crosses through zero. The forced switching is achieved by adding a controlled amount of offset to the Multiplier and Current Sense Comparator circuits. The equation shown below accounts for the built-in offsets and is accurate to within ten percent. Let Vth(M) = 1.991 V VCS, Pin 4 Threshold = 0.544 (VPin 2 - Vth(M)) VPin 3 + 0.0417 (VPin 2 - Vth(M)) Zero Current Detector The MC34262 operates as a critical conduction current mode controller, whereby output switch conduction is initiated by the Zero Current Detector and terminated when the peak inductor current reaches the threshold level established by the Multiplier output. The Zero Current Detector initiates the next on-time by setting the RS Latch at the instant the inductor current reaches zero. This critical conduction mode of operation has two significant benefits. First, since the MOSFET cannot turn-on until the inductor current reaches zero, the output rectifier reverse recovery time becomes less critical, allowing the use of an inexpensive rectifier. Second, since there are no deadtime gaps between cycles, the ac line current is continuous, thus limiting the peak switch to twice the average input current. The Zero Current Detector indirectly senses the inductor current by monitoring when the auxiliary winding voltage falls below 1.4 V. To prevent false tripping, 200 mV of hysteresis is provided. Figure 9 shows that the thresholds are well-defined over temperature. The Zero Current Detector input is internally protected by two clamps. The upper 6.7 V clamp prevents input overvoltage breakdown while the lower 0.7 V clamp prevents substrate injection. Current limit protection of the lower clamp transistor is provided in the event that the input pin is accidentally shorted to ground. The Zero Current Detector input to Drive Output turn-on propagation delay is typically 320 ns.
MOTOROLA ANALOG IC DEVICE DATA
7
MC34262 MC33262
Figure 18. Inductor Current and MOSFET Gate Voltage Waveforms
Peak
Inductor Current
Average
Timer A watchdog timer function was added to the IC to eliminate the need for an external oscillator when used in stand-alone applications. The Timer provides a means to automatically start or restart the preconverter if the Drive Output has been off for more than 620 s after the inductor current reaches zero. The restart time delay versus temperature is shown in Figure 8. Undervoltage Lockout and Quickstart An Undervoltage Lockout comparator has been incorporated to guarantee that the IC is fully functional before enabling the output stage. The positive power supply terminal (VCC) is monitored by the UVLO comparator with the upper threshold set at 13 V and the lower threshold at 8.0 V. In the stand-by mode, with VCC at 7.0 V, the required supply current is less than 0.4 mA. This large hysteresis and low startup current allow the implementation of efficient bootstrap startup techniques, making these devices ideally suited for wide input range off-line preconverter applications. An internal 36 V clamp has been added from VCC to ground to protect the IC and capacitor C4 from an overvoltage condition. This feature is desirable if external circuitry is used to delay the startup of the preconverter. The supply current, startup, and operating voltage characteristics are shown in Figures 13 and 14. A Quickstart circuit has been incorporated to optimize converter startup. During initial startup, compensation capacitor C1 will be discharged, holding the error amp output below the Multiplier threshold. This will prevent Drive Output switching and delay bootstrapping of capacitor C4 by diode D6. If Pin 2 does not reach the multiplier threshold before C4 discharges below the lower UVLO threshold, the converter will "hiccup" and experience a significant startup delay. The Quickstart circuit is designed to precharge C1 to 1.7 V, Figure 7. This level is slightly below the Pin 2 Multiplier threshold, allowing immediate Drive Output switching and bootstrap operation when C4 crosses the upper UVLO threshold. Drive Output The MC34262/MC33262 contain a single totem-pole output stage specifically designed for direct drive of power MOSFETs. The Drive Output is capable of up to 500 mA peak current with a typical rise and fall time of 50 ns with a 1.0 nF load. Additional internal circuitry has been added to keep the Drive Output in a sinking mode whenever the Undervoltage Lockout is active. This characteristic eliminates the need for an external gate pull-down resistor. The totem-pole output has been optimized to minimize cross-conduction current during high speed operation. The addition of two 10 resistors, one in series with the source output transistor and one in series with the sink output transistor, helps to reduce the cross-conduction current and radiated noise by limiting the output rise and fall time. A 16 V clamp has been incorporated into the output stage to limit the high state VOH. This prevents rupture of the MOSFET gate when VCC exceeds 20 V.
0
On MOSFET Q1 Off
Current Sense Comparator and RS Latch The Current Sense Comparator RS Latch configuration used ensures that only a single pulse appears at the Drive Output during a given cycle. The inductor current is converted to a voltage by inserting a ground-referenced sense resistor R7 in series with the source of output switch Q1. This voltage is monitored by the Current Sense Input and compared to a level derived from the Multiplier output. The peak inductor current under normal operating conditions is controlled by the threshold voltage of Pin 4 where: IL(pk ) = Pin 4 Threshold R7
Abnormal operating conditions occur during preconverter startup at extremely high line or if output voltage sensing is lost. Under these conditions, the Multiplier output and Current Sense threshold will be internally clamped to 1.5 V. Therefore, the maximum peak switch current is limited to: Ipk(max) = 1.5 V R7
An internal RC filter has been included to attenuate any high frequency noise that may be present on the current waveform. This filter helps reduce the ac line current distortion especially near the zero crossings. With the component values shown in Figure 20, the Current Sense Comparator threshold, at the peak of the haversine varies from 1.1 V at 90 Vac to 100 mV at 268 Vac. The Current Sense Input to Drive Output turn-off propagation delay is typically less than 200 ns.
8
MOTOROLA ANALOG IC DEVICE DATA
MC34262 MC33262
APPLICATIONS INFORMATION
The application circuits shown in Figures 19, 20 and 21 reveal that few external components are required for a complete power factor preconverter. Each circuit is a peak detecting current-mode boost converter that operates in critical conduction mode with a fixed on-time and variable off-time. A major benefit of critical conduction operation is that the current loop is inherently stable, thus eliminating the need for ramp compensation. The application in Figure 19 operates over an input voltage range of 90 Vac to 138 Vac and provides an output power of 80 W (230 V at 350 mA) with an associated power factor of approximately 0.998 at nominal line. Figures 20 and 21 are universal input preconverter examples that operate over a continuous input voltage range of 90 Vac to 268 Vac. Figure 20 provides an output power of 175 W (400 V at 440 mA) while Figure 21 provides 450 W (400 V at 1.125 A). Both circuits have an observed worst-case power factor of approximately 0.989. The input current and voltage waveforms of Figure 20 are shown in Figure 22 with operation at 115 Vac and 230 Vac. The data for each of the applications was generated with the test set-up shown in Figure 24.
Table 1. Design Equations
Notes Calculate the maximum required output power. Calculated at the minimum required ac line voltage for output regulation. Let the efficiency = 0.92 for low line operation. Let the switching cycle t = 40 s for universal input (85 to 265 Vac) operation and 20 s for fixed input (92 to 138 Vac, or 184 to 276 Vac) operation. In theory the on-time ton is constant. In practice ton tends to increase at the ac line zero crossings due to the charge on capacitor C5. Let Vac = Vac(LL) for initial ton and toff calculations. The off-time toff is greatest at the peak of the ac line voltage and approaches zero at the ac line zero crossings. Theta () represents the angle of the ac line voltage. The minimum switching frequency occurs at the peak of the ac line voltage. As the ac line voltage traverses from peak to zero, toff approaches zero producing an increase in switching frequency. Set the current sense threshold VCS to 1.0 V for universal input (85 Vac to 265 Vac) operation and to 0.5 V for fixed input (92 Vac to 138 Vac, or 184 Vac to 276 Vac) operation. Note that VCS must be <1.4 V. Set the multiplier input voltage VM to 3.0 V at high line. Empirically adjust VM for the lowest distortion over the ac line voltage range while guaranteeing startup at minimum line. The IIB R1 error term can be minimized with a divider current in excess of 50 A. The calculated peak-to-peak ripple must be less than 16% of the average dc output voltage to prevent false tripping of the Overvoltage Comparator. Refer to the Overvoltage Comparator text. ESR is the equivalent series resistance of C3 The bandwidth is typically set to 20 Hz. When operating at high ac line, the value of C1 may need to be increased. (See Figure 25) Calculation Required Converter Output Power Peak Inductor Current Formula PO = VO IO IL(pk) = 2 2 PO Vac(LL)
Inductance
t LP =
VO - Vac(LL) Vac(LL)2 2 2 VO PO 2 PO LP Vac2 ton VO 2 Vac Sin 1 ton + toff -1
Switch On-Time
ton =
Switch Off-Time
toff =
Switching Frequency
f=
Peak Switch Current
R7 =
VCS IL(pk)
Multiplier Input Voltage
VM =
Vac R5 R3
2 +1
Converter Output Voltage
VO = Vref
R2 +1 R1
- IIB R2
Converter Output Peak to Peak Ripple Voltage
VO(pp) = IO
1 2fac C3
2 + ESR2
Error Amplifier Bandwidth
BW =
gm 2 C1
The following converter characteristics must be chosen: Vac -- AC RMS line voltage VO -- Desired output voltage IO -- Desired output current Vac(LL) -- AC RMS low line voltage VO -- Converter output peak-to-peak ripple voltage
MOTOROLA ANALOG IC DEVICE DATA
9
MC34262 MC33262
Figure 19. 80 W Power Factor Controller
1 D2 92 to 138 RFI Vac Filter D4 Zero Current Detector + 1.2V + 36V 6.7V UVLO + 13V/ 8.0V 16V Drive Output RS Latch 2.2M R5 10 10 7 MUR130 D5 MTP 8N50E Q1 5 C5 100k R6 8 +
1N4934 D6 100 C4 22k R4 T
D1
D3
1.6V/ 1.4V
2.5V Reference
Timer R Delay
VO 230V/0.35A + 220 C3
Current Sense Comparator
20k 1.5V Overvoltage Comparator + 1.08 Vref 10A Error Amp + Vref 10pF
4 0.1 R7
1.0M R2
0.01 C2
7.5k R3
Multiplier 3
1 11k R1
Quickstart 6 2 0.68 C1
Power Factor Controller Test Data
AC Line Input Current Harmonic Distortion (% Ifund) Vrms 90 100 110 120 130 138 Pin 85.9 85.3 85.1 84.7 84.4 84.1 PF 0.999 0.999 0.998 0.998 0.997 0.996 Ifund 0.93 0.85 0.77 0.71 0.65 0.62 THD 2.6 2.3 2.2 3.0 3.9 4.6 2 0.08 0.13 0.10 0.09 0.12 0.16 3 1.6 1.0 0.58 0.73 1.7 2.4 5 0.84 1.2 1.5 1.9 2.2 2.3 7 0.95 0.73 0.59 0.58 0.61 0.60 VO(pp) 4.0 4.0 4.0 4.1 4.1 4.1 VO 230.7 230.7 230.7 230.7 230.7 230.7 IO 0.350 0.350 0.350 0.350 0.350 0.350 PO 80.8 80.8 80.8 80.8 80.8 80.8 (%) 94.0 94.7 94.9 95.3 95.7 96.0 DC Output
This data was taken with the test set-up shown in Figure 24. T = Coilcraft N2881-A Primary: 62 turns of # 22 AWG Secondary: 5 turns of # 22 AWG Core: Coilcraft PT2510, EE 25 Gap: 0.072 total for a primary inductance (LP) of 320 H Heatsink = AAVID Engineering Inc. 590302B03600, or 593002B03400
10
MOTOROLA ANALOG IC DEVICE DATA
MC34262 MC33262
Figure 20. 175 W Universal Input Power Factor Controller
1 D2 90 to 268 RFI Vac Filter D4 Zero Current Detector + 1.2V 1.6V/ 1.4V UVLO + 13V/ 8.0V 16V Drive Output RS Latch 1.3M R5 10 10 7 MUR460 D5 MTP 14N50E Q1 + 36V 6.7V 5 C5 100k R6 8 +
1N4934 D6 100 C4 22k R4 T
D1
D3
2.5V Reference
Timer R Delay
VO 400V/0.44A + 330 C3
Current Sense Comparator
20k 1.5V Overvoltage Comparator + 1.08 Vref 10A Error Amp + Vref 10pF
4 0.1 R7
1.6M R2
0.01 C2
12k R3
Multiplier 3
1 10k R1
Quickstart 6 2 0.68 C1
Power Factor Controller Test Data
AC Line Input Current Harmonic Distortion (% Ifund) Vrms 90 120 138 180 240 268 Pin 193.3 190.1 188.2 184.9 182.0 180.9 PF 0.991 0.998 0.999 0.998 0.993 0.989 Ifund 2.15 1.59 1.36 1.03 0.76 0.69 THD 2.8 1.6 1.2 2.0 4.4 5.9 2 0.18 0.10 0.12 0.10 0.09 0.10 3 2.6 1.4 1.3 0.49 1.6 2.3 5 0.55 0.23 0.65 1.2 2.3 2.9 7 1.0 0.72 0.80 0.82 0.51 0.46 VO(pp) 3.3 3.3 3.3 3.4 3.4 3.4 VO 402.1 402.1 402.1 402.1 402.1 402.1 IO 0.44 0.44 0.44 0.44 0.44 0.44 PO 176.9 176.9 176.9 176.9 176.9 176.9 (%) 91.5 93.1 94.0 95.7 97.2 97.8 DC Output
This data was taken with the test set-up shown in Figure 24. T = Coilcraft N2880-A Primary: 78 turns of # 16 AWG Secondary: 6 turns of # 18 AWG Core: Coilcraft PT4215, EE 42-15 Gap: 0.104 total for a primary inductance (LP) of 870 H Heatsink = AAVID Engineering Inc. 590302B03600
MOTOROLA ANALOG IC DEVICE DATA
11
MC34262 MC33262
Figure 21. 450 W Universal Input Power Factor Controller
2 D2 90 to 268 RFI Vac Filter D4 Zero Current Detector + 1.2V 1.6V/ 1.4V UVLO + 13V/ 8.0V 16V Drive Output RS Latch 1.3M R5 10 10 7 MUR460 D5 + MTW 20N50E Q1 + 36V 6.7V 5 C5 100k R6 8 +
1N4934 D6 100 C4 22k R4 T
D1
D3
2.5V Reference
VO 330 C3
400V/1.125A
Timer R Delay
Current Sense Comparator
20k 1.5V Overvoltage Comparator + 1.08 Vref 10A Error Amp + Vref 10pF
4
330 0.001 0.05 R7
1.6M R2
0.01 C2
12k R3
Multiplier 3
1 10k R1
Quickstart 6 2 0.68 C1
Power Factor Controller Test Data
AC Line Input Current Harmonic Distortion (% Ifund) Vrms 90 120 138 180 240 268 Pin 489.5 475.1 470.6 463.4 460.1 459.1 PF 0.990 0.998 0.998 0.998 0.996 0.995 Ifund 5.53 3.94 3.38 2.57 1.91 1.72 THD 2.2 2.5 2.1 4.1 4.8 5.8 2 0.10 0.12 0.06 0.21 0.14 0.10 3 1.5 0.29 0.70 2.0 4.3 5.0 5 0.25 0.62 1.1 1.6 2.2 2.5 7 0.83 0.52 0.41 0.71 0.63 0.61 VO(pp) 8.8 8.8 8.8 8.9 8.9 8.9 VO 395.5 395.5 395.5 395.5 395.5 395.5 IO 1.14 1.14 1.14 1.14 1.14 1.14 PO 450.9 450.9 450.9 450.9 450.9 450.9 (%) 92.1 94.9 95.8 97.3 98.0 98.2 DC Output
This data was taken with the test set-up shown in Figure 24. T = Coilcraft P3657-A Primary: 38 turns Litz wire, 1300 strands of #48 AWG, Kerrigan-Lewis, Chicago, IL Secondary: 3 turns of # 20 AWG Core: Coilcraft PT4220, EE 42-20 Gap: 0.180 total for a primary inductance (LP) of 190 H Heatsink = AAVID Engineering Inc. 604953B04000 Extrusion
12
MOTOROLA ANALOG IC DEVICE DATA
MC34262 MC33262
Figure 22. Power Factor Corrected Input Waveforms (Figure 20 Circuit)
Input = 115 VAC, Output = 175 W Input = 230 VAC, Output = 175 W
Voltage = 100 V/DIV
Voltage = 100 V/DIV
Current = 1.0 A/DIV
2.0 ms/DIV
2.0 ms/DIV
Figure 23. Output Voltage Startup Overshoot (Figure 20 Circuit)
Without Overvoltage Comparator 500 V 26% 400 V 432 V 400 V With Overvoltage Comparator 8%
80 V/DIV
0V 200 ms/DIV
0V 200 ms/DIV
Figure 24. Power Factor Test Set-Up
2X Step-Up Isolation Transformer
RFI Test Filter AC POWER ANALYZER PM 1000
W VA PF Vrms Arms
Line
115 Vac Input Neutral
HI
HI
T
Autoformer I O
A
LO LO
V
0.1
0.005 1.0 0.005
0
Vcf 7
1
9
2
11
3
13
5 Voltech
0 to 270 Vac Output to Power Factor Controller Circuit
Acf Ainst FREQ HARM
Earth
An RFI filter is required for best performance when connecting the preconverter directly to the ac line. The filter attenuates the level of high frequency switching that appears on the ac line current waveform. Figures 19 and 20 work well with commercially available two stage filters such as the Delta Electronics 03DPCG5. Shown above is a single stage test filter that can easily be constructed with four ac line rated capacitors and a common-mode transformer. Coilcraft CMT3-28-2 was used to test Figures 19 and 20. It has a minimum inductance of 28 mH and a maximum current rating of 2.0 A. Coilcraft CMT4-17-9 was used to test Figure 21. It has a minimum inductance of 17 mH and a maximum current rating of 9.0 A. Circuit conversion efficiency (%) was calculated without the power loss of the RFI filter.
MOTOROLA ANALOG IC DEVICE DATA
13
80 V/DIV
Current = 1.0 A/DIV
MC34262 MC33262
Figure 25. Error Amp Compensation
10A
Error Amp + 1
R2
R1 6 C1 2
The Error Amp output is a high impedance node and is susceptible to noise pickup. To minimize pickup, compensation capacitor C1 must be connected as close to Pin 2 as possible with a short, heavy ground returning directly to Pin 6. When operating at high ac line, the voltage at Pin 2 may approach the lower threshold of the Multiplier, 2.0 V. If there is excessive ripple on Pin 2, the Multiplier will be driven into cut-off causing circuit instability, high distortion and poor power factor. This problem can be eliminated by increasing the value of C1.
Figure 26. Current Waveform Spike Suppression
Figure 27. Negative Current Waveform Spike Suppression
7
7
22k 10pF Current Sense Comparator
4 C
R R7 10pF Current Sense Comparator
22k
4 D1 R7
A narrow turn-on spike is usually present on the leading edge of the current waveform and can cause circuit instability. The MC34262 provides an internal RC filter with a time constant of 220 ns. An additional external RC filter may be required in universal input applications that are above 200 W. It is suggested that the external filter be placed directly at the Current Sense Input and have a time constant that approximates the spike duration.
A negative turn-off spike can be observed on the trailing edge of the current waveform. This spike is due to the parasitic inductance of resistor R7, and if it is excessive, it can cause circuit instability. The addition of Shottky diode D1 can effectively clamp the negative spike. The addition of the external RC filter shown in Figure 26 may provide sufficient spike attenuation.
14
MOTOROLA ANALOG IC DEVICE DATA
MC34262 MC33262
Figure 28. Printed Circuit Board and Component Layout (Circuits of Figures 15 and 16)
(Top View)
3.0
4.5 (Bottom View)
NOTE: Use 2 oz. copper laminate for optimum circuit performance.
MOTOROLA ANALOG IC DEVICE DATA
15
MC34262 MC33262
OUTLINE DIMENSIONS
P SUFFIX PLASTIC PACKAGE CASE 626-05 ISSUE K -B-
1 4
8
5
NOTES: 1. DIMENSION L TO CENTER OF LEAD WHEN FORMED PARALLEL. 2. PACKAGE CONTOUR OPTIONAL (ROUND OR SQUARE CORNERS). 3. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. DIM A B C D F G H J K L M N MILLIMETERS MIN MAX 9.40 10.16 6.10 6.60 3.94 4.45 0.38 0.51 1.02 1.78 2.54 BSC 0.76 1.27 0.20 0.30 2.92 3.43 7.62 BSC --- 10_ 0.76 1.01 INCHES MIN MAX 0.370 0.400 0.240 0.260 0.155 0.175 0.015 0.020 0.040 0.070 0.100 BSC 0.030 0.050 0.008 0.012 0.115 0.135 0.300 BSC --- 10_ 0.030 0.040
F
NOTE 2
-A- L
C -T-
SEATING PLANE
J N D K
M
M
H
G 0.13 (0.005) TA
M
B
M
-A-
8 5
D SUFFIX PLASTIC PACKAGE CASE 751-05 (SO-8) ISSUE N
-B-
1 4
4X
P 0.25 (0.010)
M
B
M
NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: MILLIMETER. 3. DIMENSIONS A AND B DO NOT INCLUDE MOLD PROTRUSION. 4. MAXIMUM MOLD PROTRUSION 0.15 (0.006) PER SIDE. 5. DIMENSION D DOES NOT INCLUDE DAMBAR PROTRUSION. ALLOWABLE DAMBAR PROTRUSION SHALL BE 0.127 (0.005) TOTAL IN EXCESS OF THE D DIMENSION AT MAXIMUM MATERIAL CONDITION. MILLIMETERS MIN MAX 4.80 5.00 3.80 4.00 1.35 1.75 0.35 0.49 0.40 1.25 1.27 BSC 0.18 0.25 0.10 0.25 0_ 7_ 5.80 6.20 0.25 0.50 INCHES MIN MAX 0.189 0.196 0.150 0.157 0.054 0.068 0.014 0.019 0.016 0.049 0.050 BSC 0.007 0.009 0.004 0.009 0_ 7_ 0.229 0.244 0.010 0.019
G C -T-
8X SEATING PLANE
R
X 45 _
F
D 0.25 (0.010)
M
K TB
M_
S
J
S
A
DIM A B C D F G J K M P R
Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation consequential or incidental damages. "Typical" parameters which may be provided in Motorola data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including "Typicals" must be validated for each customer application by customer's technical experts. Motorola does not convey any license under its patent rights nor the rights of others. Motorola products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the Motorola product could create a situation where personal injury or death may occur. Should Buyer purchase or use Motorola products for any such unintended or unauthorized application, Buyer shall indemnify and hold Motorola and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that Motorola was negligent regarding the design or manufacture of the part. Motorola and are registered trademarks of Motorola, Inc. Motorola, Inc. is an Equal Opportunity/Affirmative Action Employer. How to reach us: USA / EUROPE / Locations Not Listed: Motorola Literature Distribution; P.O. Box 20912; Phoenix, Arizona 85036. 1-800-441-2447 or 602-303-5454 MFAX: RMFAX0@email.sps.mot.com - TOUCHTONE 602-244-6609 INTERNET: http://Design-NET.com
JAPAN: Nippon Motorola Ltd.; Tatsumi-SPD-JLDC, 6F Seibu-Butsuryu-Center, 3-14-2 Tatsumi Koto-Ku, Tokyo 135, Japan. 03-81-3521-8315 ASIA/PACIFIC: Motorola Semiconductors H.K. Ltd.; 8B Tai Ping Industrial Park, 51 Ting Kok Road, Tai Po, N.T., Hong Kong. 852-26629298
16
*MC34262/D*
MOTOROLA ANALOG IC DEVICE DATA MC34262/D

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